Dielectric resonator tuning device

ABSTRACT

A temperature-compensating tuning device is disclosed for tuning and temperature stabilizing the resonant frequency of a dielectric resonator. The tuning device comprises a tuning element in the form of a cylindrical shaft, an inner sleeve coaxially around the tuning element and mating therewith by corresponding sets of threads, and an outer sleeve coaxially around the inner sleeve, and mating therewith by corresponding sets of threads. The outer surface of the outer sleeve may also include threads for mating with threads of a dielectric resonator enclosure. Rotation of the tuning element, inner sleeve and/or outer sleeve can move the tuning element in proximity to a dielectric resonator, which provides the resonant frequency tuning effect. The tuning element, inner and outer sleeves are made of temperature expanding material to cause the tuning element to move in proximity of the dielectric resonator with temperature changes to provide temperature stability to the resonant frequency.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation-in-part of patent application Ser.No. 09/082,805 filed on May 21, 1998, which is incorporated herein byreference for all purposes. In addition, this application is related toapplication entitled “RF/Microwave Oscillator” filed on even date, whichis incorporated herein by reference for all purposes.

FIELD OF THE INVENTION

This invention relates generally to radio frequency or microwavecircuits, and specifically to techniques for improving the directcurrent (DC) biasing of oscillators, amplifiers, and other RF/Microwavedevices; for improving the gate return for a field effect transistorused in an oscillator; for providing a cavity or enclosure for adielectric resonator that provides improved temperature stability over awide frequency range; for providing an improved microwave oscillator;and for providing an improved dielectric resonator tuning device.

BACKGROUND OF THE INVENTION

Frequency-Adjustable DC Biasing Circuit

Dielectric resonator oscillators (DROs) are very popular devices in theradio frequency (RF) or microwave electronic field. These oscillatorsare typically employed in communication systems, radar systems,navigation systems and other signal receiving and/or transmittingsystems. Their popularity has been attributed to their high-Q, low loss,and conveniently sized devices for various applications in the RF andmicrowave fields. For the purpose of this application, the terms “radiofrequency”, “RF” and “microwave” are interchangeable, and are used torefer to the field of electronics that involve signal processing ofelectromagnetic energy cycling at a frequency range of about 800 MHz toabout 300 GHz.

Although DROs provide substantial advantages over other types ofoscillator designs, improving their performance and characteristics isan ongoing process. For instance, some ongoing developments includereducing the size of the DROs, increasing its efficiency, improving itsmanufacturing and reliability, reducing its phase noise, and improvingits temperature stability. Of particular interest to this invention isthe latter three objectives.

Manufacturers of DROs are concerned with improving the manufacturing andreliability of their products. The design of DROs presents a particularproblem in that DROs typically perform well only for an RF energy orsignal cycling at a discreet frequency or within a narrow frequencyrange. In other words, they generally meet their specified performanceonly for a very narrow frequency range. It follows then that if a DROmanufacturer wants to produce a line of DROs with different discreetoutput frequencies covering a wide frequency range, each DRO must becustom tailored for each of the frequencies. This custom tailoring ofDROs leads to increased engineering time, manufacturing time, cost,inventory and logistics, and a reduction in the reliability of the DROs.

To illustrate the manufacturing and reliability problem of customizingDROs, consider the typical prior art series feedback or reflective typeDRO 10 shown in FIG. 1. The DRO 10 consists of a dielectric resonator(DR) 12, field effect transistor (FET) 14, a DR-coupling or inputresonator transmission line 16, output and source impedance matchingcircuits 18 and 20, direct current (DC) biasing circuits 22 and 24 forthe drain and source of the FET, and a FET gate return resistor R3.

In the prior art, the DRO 10 is typically designed for efficiently andoptimally producing an RF energy or signal cycling at one specificfrequency, or within a very narrow frequency range. For example, the DRO10 is specifically designed to produce an RF energy or signal cycling ata frequency f₀. In order for the DRO 10 to optimally perform, each ofthe elements of the DRO is tailored designed to optimally operate atsuch frequency f₀. For instance, the dielectric resonator 12 is chosensuch that its lowest resonating frequency is slightly below thefrequency f₀. Similarly, the output and source impedance matchingcircuits 18 and 20 are designed to provide the optimal impedancematching at frequency f₀. Also, the drain and source DC biasing circuits22 and 24 are designed so that they optimally block an RF energy orsignal cycling at the operating frequency of the DRO f₀.

To further illustrate the need for optimally designing each of theelements of the DRO 10 for its operating frequency f₀, consider forexample the drain and source DC biasing circuits 22 and 24. The objectof these circuits is to transmit DC power to the FET 14 withoutaffecting the RF energy produced by the DRO 10. To accomplish thisobjective, the source and drain DC biasing circuits 22 and 24 includerespective high impedance transmission lines 26 and 28, each having oneend (RF end) connected to an RF-carrying portion of the DRO 10, andanother end (DC end) being RF shunted to ground by a bypass capacitor,such as capacitors C1 and C2.

In order for the DC biasing circuits to optimally not affect the RFenergy or signal produced by the DRO 10, the length of the highimpedance transmission lines 26 and 28 are designed to have a length ofa quarter wavelength of an RF energy cycling at the operating frequencyof the DRO f₀. In addition, the bypass capacitors C1 and C2 are designedto produce an impedance to ground of less than one to two Ohms at thefrequency f₀. The biasing circuits 22 and 24 typically include resistorsR1 and R2 for setting the proper bias voltage for the FET 14. Anydeviation of the length of the high impedance transmission lines 26 and28 from a quarter wavelength length at the operating frequency f₀ of theDRO 10 will cause degradation in the performance of the DRO, such as adegradation in the phase noise performance of the device.

From the discussion above, it can be seen that in the prior art DRO 10,the elements of the DRO 10 are tailored designed for optimally operatingat the specific operating frequency f₀ of the DRO. This presents aproblem for manufactures of DROs that need to produce a line of DROsoperating at a plurality of different discreet frequencies covering awide frequency range. In other words, because each type of DROs must becustom designed, it leads to increased engineering time, manufacturingtime, cost, inventory and logistics, and a reduction in the reliabilityof the DROs. Thus, there is a need for a universal DRO design that canbe easily modified to optimally operate at a plurality of differentdiscreet frequencies covering a wide frequency range.

Improved FET Gate Return Circuit

Another concern in the design of DROs is the phase noise performance ofthe device. Reduction in phase noise is desired since high phase noisemay affect the performance of systems employing DROs. For instance, DROsoften produce an RF carrier that is to be modulated with a basebandsignal. If the frequency response of the baseband signal include arelatively low frequency response, its frequency components lie near theRF carrier. If the RF carrier has poor phase noise characteristics, thenit will interfere with the modulated baseband signal. Thus, it isdesired to reduce phase noise as much as possible in DROs to avoid thisinterference problem.

Referring again to FIG. 1, one particular element of the prior art DRO10, namely the FET gate return resistor R3, can cause significantdegradation in the phase noise performance of the DRO. The gate returnresistor R3 is typically connected in a series feedback or reflectivetype DRO at the end of the DR-coupling or input resonator transmissionline 16. The purpose of the gate return resistor R3 is to provide a pathto ground for positive charges that accumulate on the gate of the FET 14during its operation.

More specifically, during the operation of the DRO 10, a large signalamplitude is generated at the gate input of the FET 14. As the largesignal amplitude varies over the positive half of the sinusoid wave, asmall amount of positive charges pass through the Schottky diodejunction of the gate. These charges interfere with operation of the DRO,and therefore, need to be removed. Thus, the FET gate resistor R3provides a path to ground to eliminate such unwanted charges. In orderto eliminate any unwanted RF reflections off the FET gate resistor R3,this resistor is designed to match the characteristic impedance of theDR-coupling or resonator transmission line 16, which is typically 50Ohms.

Although the problem of the unwanted positive charges at the input ofthe DRO 10 is substantially reduced by the FET gate resistor R3, thisresistor has an adverse effect of degrading the phase noise performanceof the DRO. The reasons for this is that the resistance value of theresistor R3 is relatively low, e.g. 50 Ohms, and it is not properly RFisolated from the DRO circuitry, i.e., it is directly connected to theend of the DR-coupling or resonator transmission line 16. As a result,the resistor affects the RF energy propagating via that DR-coupling orresonator transmission line 16, and consequently, adversely affects thephase noise of the DRO. Accordingly, there is a need to provide a FETgate return resistor that provides a path to ground for the unwantedpositive charges emanating from the FET 14, without significantlydegrading the phase noise performance of the DRO.

Dielectric Resonator Cavity

Yet another concern in the design of DROs is the temperature stabilityof the device. Although DROs have superior performance when it comes tophase noise and efficiency, DROs are susceptible to environmenttemperature changes if they are not properly designed. Therefore, agreat deal of engineering time is spent in designingtemperature-compensating elements and/or techniques for DROs.

For instance, in FIG. 2, a prior art temperature-compensating DROcircuit 30 is shown. The circuit 30 includes a DRO, such as like theprior art DRO 10 of FIG. 1, coupled to a phase lock loop (PLL) circuit32. The PLL circuit 32 includes a crystal oscillator 34, a samplingphase detector 36 and a loop filter 38. As it is conventionally known,the crystal oscillator 34 produces a highly temperature-stablesinusoidal signal with typically a much lower frequency f_(x) than thefrequency f₀ of the DRO output. This sinusoidal signal is coupled to afirst input of the sampling phase detector 36, whereas the outputsinusoidal signal of the DRO 10 is coupled to a second input of thesampling phase detector 36 by way of a directional coupler 40. Thesampling phase detector 36 produces a phase error signal which iscoupled to a frequency-responsive component (not shown), such as avaractor, of the DRO 10 by way of a loop filter 38.

Since the output of the crystal oscillator 34 is highly temperaturestable, the output of the DRO 10 is also highly stable since the PLLcircuit causes the stability of the DRO output frequency f₀ to track thestability of the frequency f_(x) of the crystal oscillator 34. Hence,with the PLL circuit 32, the DRO 10 is temperature stable, or as good asthe temperature stability of the crystal oscillator 34.

However, this temperature stability does not come without a price. Forinstance, the temperature compensated DRO circuit includes additionalcomponents, such as the crystal oscillator 34, sampling phase detector36, loop filter 38, varactor (not shown) and directional coupler 40.These additional elements add to the costs of the DRO, increases theengineering and manufacturing time, increases inventory, complicateslogistics, and reduces the reliability of the DRO circuit. Thus, thereis a need for a temperature-compensated DRO that does not require suchadditional elements. In addition, there is a further need to providesuch temperature compensation in a manner that applies to a plurality ofoperating frequencies so that the DROs need not be custom made.

Temperature-Compensating Resonant Frequency Tuning Device

The reason a DRO or any other dielectric resonator apparatus requiretemperature compensating circuitry is that the resonant frequency of adielectric resonator can vary as a function of temperature. In moredetail, the resonant frequency of a dielectric resonator is a functionof its geometry and size. Typically, standard dielectric resonatorsavailable in the market are of cylindrical shape, and are often referredto as dielectric resonator pucks. Therefore, for a dielectric resonatorpuck, the resonant frequency is dependent on the puck's diameter (Dr)and height (Lr). More accurately, the resonant frequency variesinversely with the puck's diameter (Dr) and height (Lr), i.e. the largerthe puck's diameter (Dr) and height (Lr) are, the smaller the resonantfrequency is, and vice-versa.

Dielectric resonators are typically comprised of temperature-expandingmaterials. Accordingly, as the environment temperature changes, thediameter (Dr) and height (Lr) of the dielectric resonator also changewith temperature. Generally, as temperature rises, the dielectricresonator material expands, causing its diameter (Dr) and height (Lr) toincrease. Conversely, as temperature drops, the dielectric resonatormaterial contracts, causing its diameter (Dr) and height (Lr) todecrease. The resonant frequency of a dielectric resonator, on the otherhand, varies inversely with the dielectric resonator size. That is, thelarger the resonator's diameter (Dr) and height (Lr) are, the smallerits resonant frequency is, and vice-versa.

It follows then that the resonant frequency of a dielectric resonatorvaries inversely with temperature variation. That is, as temperaturerises, the resonant frequency of the dielectric resonator decreases, andas temperature drops, the resonant frequency increases. Because of thedependent relationship between the resonant frequency of a dielectricresonator and the environment temperature, achieving a desiredtemperature stability for dielectric resonator devices and/or circuitsis not an easy task. As previously discussed, temperature compensatingtechniques, like the PLL technique discussed above, have been developedbut are generally expensive and require sophisticated circuitry.

FIG. 3 shows a cross-sectional view of a prior art,temperature-uncompensated, dielectric resonator apparatus 50 thatincludes a resonant frequency tuning device 60. The dielectric resonatorapparatus 50, which can include filters and dielectric resonatoroscillators (DROs), consists of a metallized housing 52 configured toform an enclosed cavity 54. A dielectric resonator puck 56 mounted on astand-off 58 is situated near the bottom of the housing 52 within thecavity 54. For resonant frequency tuning purposes, a tuning screw 60 isrotatably mounted through the top of the housing 52 and is held insecured place by a hex nut 62.

The tuning device 60 works on the principle that a metal object inproximity to a dielectric resonator affects or alters the resonantfrequency of the dielectric resonator. More specifically, as a metalobject approaches in proximity to a dielectric resonator, it interactswith the electromagnetic field present around the resonator, causing theresonant frequency to increase. As the metal object is removed, itinteracts less with the resonator's electromagnetic fields, causing theresonant frequency to decrease, until it is no longer affected by themetal object.

The tuning device 60 is such a metal object that can be positioned inproximity to the dielectric resonator to control or tune its resonantfrequency. Typically, these prior art tuning device are configured intoa threaded screw and mounted through the top of the cavity directlyabove the dielectric resonator 56. By rotating the tuning screw 60, theend of the screw can be positioned near the dielectric resonator tocause its resonant frequency to shift. In this manner, the screw can bepositioned in order to obtain the desired resonant frequency for thedielectric resonator 60. Generally, the dielectric resonator is chosenso that its fundamental or unaffected resonant frequency is slightlylower than the desired resonant frequency. The end of the tuning screw60 is then brought near the resonator to shift its resonant frequency upto the desired frequency.

The dielectric resonator apparatus 50 as shown in FIG. 3 does notinclude any temperature compensating device and/or circuits forstabilizing the resonant frequency with changes in temperature. Sincethe tuning screw 60 can alter the resonant frequency of the dielectricresonator, it would be highly desired if such a screw can be configuredto counteract the shifts in the resonant frequency due to temperaturechanges. In other words, it would be highly desirable for a tuningdevice or screw that can be used not only for setting the desiredresonant frequency of the dielectric resonator, but also to providestability of the resonant frequency during temperature variation.

Puckless RF/Microwave Oscillator

As discussed previously, temperature compensating dielectric resonatordevices and/or circuits is not an easy task. It involves substantialengineering time to properly design, manufacturing time to reliablybuild, and technician time to tune and test. Not only that, it requiressubstantially more inventory, cost and logistics for the extracomponents needed to provide the required temperature compensation. Inother words, although these dielectric resonator devices and/or circuitsachieve superior performance because of the high-Q and low lossproperties, temperature compensating these devices make them not asattractive and desirable.

For instance, it would be highly desirable for an RF/Microwaveoscillator that could achieve the desired performances of a dielectricresonator oscillator (DRO), such as high-Q, low-loss, and low phasenoise attributes, without a dielectric resonator or a resonator that issubstantially temperature dependent. Such an oscillator would not needthe complicated temperature compensating techniques needed for DROs. Ifthe temperature compensating components were to be eliminated, thiswould substantially reduce engineering time, manufacturing time andtechnician time. It would also provide substantial savings in costbecause of less components, inventory and logistics. Furthermore, suchan oscillator would be more reliable because of its fewer parts.Accordingly, there is a need for an RF/Microwave oscillator thatachieves performances comparable to a DRO, without requiring adielectric resonator.

OBJECTS OF THE INVENTION

The following includes some, but not all, of the objects achieved by thedisclosed invention:

It is a general object of the invention to provide a new and improveddielectric resonator oscillator (DRO);

It is an object of the invention to provide a DRO that can be easilymodified to optimally operate at a plurality of different discreetfrequencies covering a wide frequency range;

It is another general object of the invention to provide a new andimproved amplifier;

It is another object of the invention to provide an RF amplifier thatcan be easily modified to optimally operate at different frequencyranges;

It is another general object of the invention to provide a new andimproved DC biasing or grounding circuit for an RF circuit;

It is another object of the invention to provide a DC biasing orgrounding circuit that is easily tunable for a plurality of differentdiscreet frequencies;

It is another object of the invention to provide such easily tunable DCbiasing or grounding circuit for a DRO;

It is another object of the invention to provide such easily tunable DCbiasing or grounding circuit for an RF amplifier;

It is another general object of the invention to provide a cavity orhousing for a dielectric resonator (DR);

It is another object of the invention to provide a cavity or housingthat provides improved temperature stability for a DR device;

It is another object of the invention to provide a cavity or housingthat has improved temperature stability for a DR device capable ofoperating at a plurality of different discreet frequencies covering awide frequency range;

It is another object of the invention to provide a cavity or housing fora DRO;

It is another object of the invention to provide a cavity or housing fora dielectric resonator filter;

It is a general object of the invention to provide an improved tuningdevice for tuning the resonant frequency of a dielectric resonator;

It is an object of the invention to provide a tuning device that can beeasily adjusted to provide accurate tuning of the resonant frequency ofa dielectric resonator;

It is another object of the invention to provide a tuning device thatcan substantially temperature stabilize the resonant frequency of adielectric resonator;

It is also another object of the invention to provide a tuning devicethat includes multiple settings and/or adjustments so that the desiredtemperature stability of the resonant frequency of a dielectricresonator can be achieved;

It is another object of the invention to provide a dielectric resonatorapparatus that uses the tuning device of the invention;

It is another object of the invention to provide a dielectric resonatoroscillator that uses the tuning device of the invention;

It is yet another object of the invention to provide a dielectricresonator filter that uses the tuning device of the invention;

It is another general object of the invention to provide an improvedRF/Microwave oscillator;

It is an object of the invention to provide an RF/Microwave oscillatorthat has the quality factor (Q) performance, low-loss, and phase noisecharacteristic comparable to a dielectric resonator oscillator; and

It is an object of the invention to provide an RF/Microwave oscillatorthat has the quality factor (Q) performance, low-loss, and phase noisecharacteristics comparable to a dielectric resonator oscillator, butwith improved temperature stability.

SUMMARY OF THE INVENTION

A first aspect of the invention includes a frequency-adjustable directcurrent (DC) biasing or grounding circuit for any radio frequency (RF)circuit that requires a biasing or grounding circuit, such as adielectric resonator oscillator (DRO), an RF amplifier, a mixer, a pinattenuator, and a multiplier. The advantage of having afrequency-adjustable (DC) biasing or grounding circuit is that a singledesign can be used for numerous RF circuits that have differentfrequency responses. The frequency-adjustable biasing or groundingcircuit merely requires minimal tuning so that it can best operate at adesired frequency or frequency range.

The DC biasing or grounding circuit of the invention preferably includesa transmission line for propagating therethrough a direct current or alow frequency signal, and for substantially blocking or isolating an RFenergy or signal cycling at a selected frequency or frequency range. Thetransmission line includes a first portion thereof, preferably an end(referred to herein as an RF end), for connection to an RF circuit. Thetransmission line includes a second portion thereof, preferably anopposite end (referred to herein as a DC end), for connection to a biasvoltage, direct path to ground or a direct path to ground by way of aresistor. The electrical length between the first and second portions ofthe transmission line is preferably about 90 degrees or about an oddmultiple thereof (i.e. 270, 450, 630 . . . Etc. degrees) for an RFenergy or signal cycling at a pre-determined frequency, preferably theupper frequency in a prescribed frequency range.

The DC biasing or grounding circuit of the invention includes a lowimpedance structure to RF ground coupled to the transmission line atabout the second portion thereof, or alternatively, the DC end of thetransmission line. Because the electrical length between the first andsecond portions of the transmission line is about 90 degrees or about anodd multiple, the low impedance structure at the second portion (e.g. DCend) translates into a high impedance, or preferably a substantiallymaximized normalized impedance, at the first portion (e.g. RF end) ofthe transmission line for an RF energy or signal cycling at thepre-determined higher frequency.

In order to make the DC biasing or grounding circuit of the inventionadjustable such that it can present a larger impedance, or preferably asubstantially maximized normalized impedance, for RF energies or signalscycling at frequencies other than the pre-determined frequency, thecircuit includes a tuning element coupled to the transmission line atabout its first portion (e.g. RF end). Preferably, the tuning element isan open ended transmission line that has a length that can be adjusted.By adjusting the length of the open ended transmission line to aparticular length, the DC biasing or grounding circuit can present ahigher impedance, or preferably a substantially maximized normalizedimpedance, for an RF energy or signal cycling at a correspondingselected frequency.

Another aspect of the invention is an oscillator, preferably a DRO, thatuses the frequency-adjustable DC biasing circuit described above toprovide a bias voltage for its active device, such as a field effecttransistor, bipolar junction transistor or the like. Thefrequency-adjustable DC biasing is particularly useful for a line ofDROs producing different discrete output frequencies within a specifiedfrequency range. The advantage of the frequency-adjustable DC biasingcircuit is that a single design thereof can be incorporated into any ofa number of DROs producing different discreet frequency signals.

In the preferred embodiment, the DRO includes an active device, such asa field effect transistor (FET); a dielectric resonator coupled to aport of the active device, such as the gate of the FET; and anadjustable-frequency biasing circuit for biasing the active device, suchas the FET. The DRO may include impedance matching circuits asappropriate. With the frequency-adjustable DC biasing circuit, modifyingthe DRO for a different output frequency is simply done by adjusting theDC biasing circuit so that it provides substantially optimized RFblockage at the specified frequency, providing the proper dielectricresonator puck for the specified frequency, and performing minor tuningon the impedance matching circuits if appropriate to do so. With aversatile DC biasing circuit, a single design can be used on a pluralityof different DROs. This leads to reduced costs, manufacturing andengineering efforts, inventory, logistics, and an improvement in thereliability of the DROs.

Another aspect of the invention is an improved technique for providing agate return for a field effect transistor (FET) used in oscillators thatleads to reduced phase noise. The FET gate return of the inventionincludes a relatively high resistor, for example of about at least 10kilo Ohms. The resistor is connected to ground and coupled to the DRO byway of a high characteristic impedance transmission line, such that anend or portion thereof is coupled to the FET, preferably near the gate.The high impedance transmission line includes a length of about 90electrical degrees (quarter wavelength) or about an odd multiple thereofat the operating frequency of the DRO. Preferably, the highcharacteristic impedance transmission line is connected to the inputresonator transmission line if a series-feedback DRO is used. An RFbypass capacitor is preferably connected across the gate returnresistor.

Because the FET gate resistor is of relatively high resistance, forexample of about at least 10 kilo Ohms, coupled with the fact that it isfurther RF isolated from the DRO by the quarterwave transmission line,the DRO acquires improved phase noise performance. The FET gate returncircuit of the invention can also be made frequency-adjustable, similarto the biasing or grounding circuit discussed above, such that asubstantially maximized RF isolation is provided for other differentoutput frequencies of the DRO. This leads to improved phase noiseperformance for the DRO at the selected frequency.

Yet another aspect of the invention is a cavity or enclosure for adielectric resonator that provides improved temperature stability over awide range of frequencies. This cavity would be particularly useful fora line of DROs that output different frequency signals, wherein a singlecavity design could be used for all of the DROs and provide the neededtemperature stability.

In particular, the dielectric resonator cavity of the invention includesa width or diameter Dc and a height Lc. It is designed to house adielectric resonator structure, such as a dielectric resonator puck,having a width or diameter of Dr and a height Lr. According to theinvention, in order to provide sufficient temperature stability, it ispreferred that the diameter Dc of a cylindrical cavity be at least about3 to about 7.5 times the width or diameter Dr of the enclosed dielectricresonator, and the height Lc of the cavity be at least about 3 to about7.5 times the height Lr of the enclosed dielectric resonator. For asquare cavity, it is preferred that the width Dc be at least about 3 toabout 7.5/{square root over (2)} times the width or diameter Dr of theenclosed resonator, and the height Lc be at least about 3 to about7.5/{square root over (2)} times the height Lr of the encloseddielectric resonator. Since the resonant frequency of standarddielectric resonator puck linearly correlates with the diameter of theresonator, the cavity provides for improved temperature stability for afrequency range of more than an octave.

Because the dielectric resonator cavity of the invention can accommodatedielectric resonators having resonant frequencies that can differ bymore than an octave, a single cavity design can be used on a line ofDROs producing outputs that fall within the working frequency range ofthe cavity. This is of particular advantage since a single cavity designwould facilitate manufacturing and engineering efforts, reduce costs,inventory and logistics, and improve the reliability of the DROs. Notonly would it be useful for a line of DROs, but it would also be usefulfor a line of dielectric resonator filters having different frequencyresponses falling within the working range of the cavity. In addition,the cavity could also be used for other dielectric resonatorapplications.

In another aspect of the invention, a dielectric resonator tuning deviceis provided herein that is capable of providing substantial temperaturestability to the resonant frequency of a dielectric resonator. Or, inother words, the tuning device has temperature compensation capabilityfor substantially stabilizing the resonant frequency of a dielectricresonator as temperature varies. The temperature compensation feature ofthe tuning device of the invention works on the principle that as ametal object approaches a dielectric resonator, its resonant frequencytends to increase. Conversely, as a metal object is removed from theproximity of the dielectric resonator, its resonant frequency tends todecrease until it reaches its unaffected or fundamental resonantfrequency.

More specifically, the tuning device can be configured such that whentemperature increases, a tuning element approaches the dielectricresonator tending to cause its resonant frequency to increase. Thisaction counteracts the tendency of the resonant frequency to decrease asthe dielectric resonator expands due to the increasing temperature.Conversely, the tuning device can be configured such that whentemperature decreases, a tuning element moves away from the dielectricresonator tending to cause its resonant frequency to decrease. Thisaction counteracts the tendency of the resonant frequency to increase asthe dielectric resonator contracts due to the decreasing temperature.

One of the advantages of the tuning device of the invention is that itcan be configured to provide a large range of movement with temperaturevariation. The reason for this is that the tuning device of theinvention includes at least three materials that contribute to themovement of the tuning element with temperature variations. Thecompositions of these materials can be chosen so that the cumulativeexpansion/contraction of these materials with temperature variationprovides the needed movement of the tuning element to obtain the desiredtemperature stability of the dielectric resonator. Another advantage ofthe tuning device of the invention is that the individual contributionsof the materials to the movement of the tuning element can be easilyadjusted to precisely achieve the needed movement of the tuning elementto provide the desired temperature stability of the resonant frequencyof the dielectric resonator.

In more detail, the dielectric resonator tuning device is preferablycomprised of a tuning element, preferably in the form of an elongatedcylindrical shaft, which can be solid or hollow. The tuning elementincludes a portion, preferably an end of the shaft, forelectromagnetically interacting with a dielectric resonator. The otherend of the shaft is attached to a head portion which includes a slot forreceiving a mechanical device, like a screw driver, to assist in therotation of the tuning element. An inner sleeve is situated coaxiallyaround the tuning element and head portion, and includes an inner set ofthreads configured to mate with a set of threads formed on the sides ofthe tuning element and on the side of the head portion. An outer sleeveis situated coaxially around the inner sleeve and includes an inner setof threads configured to mate with an outer set of threads of the innersleeve. The outer sleeve also includes an outer set of threads forrotational attachment to a dielectric resonator housing or an additionalsleeve, if needed.

The linear temperature expansion and contraction of the tuning element,the inner sleeve and the outer sleeve contribute to the overall movementof the tuning element with temperature variations. The desired range ofmovement of the tuning element can be achieved by proper selection ofthe materials for the tuning element and inner and outer sleeves. Also,the movement of the tuning element is a function of the length of thetuning element and inner and outer sleeves below their respective pointof contact to their adjacent, outer element. These lengths can bechanged by rotation of these elements for providing the desired movementof the tuning element.

In another aspect of the invention, an RF/Microwave oscillator isprovided herein that is characterized by having high-Q, low-loss, andlow phase noise performance, comparable to a dielectric resonatoroscillator, with the added benefit of having a resonant circuit that issubstantially invariant with changes in temperature. In the preferredembodiment, the RF/Microwave oscillator includes an active device,preferably a field effect transistor or the like, that has threeterminals, such as a gate, source and drain, coupled to a tune line orresonator circuit, a feedback circuit preferably of a series type, andan output circuit, respectively. Each of these circuits comprises atleast a pair of coupled transmission lines, preferably formed on asubstrate in a microstrip form, and is designed to resonatesubstantially at the operating frequency of the oscillator.

The high-Q, low-loss, and phase noise performance of the RF/Microwaveoscillator can be improved by including in the resonator, feedback andoutput circuits, multiple pairs of coupled transmission lines beingcascaded together. The high-Q and low-loss properties of theRF/Microwave oscillator are proportional to the number of cascaded pairsof coupled transmission lines present in the resonator, feedback andoutput circuits. By including a sufficient amount of cascaded coupledtransmission lines in these circuits, the high-Q and low-loss propertiesof a DRO can be achieved. The advantage of the RF/Microwave oscillatorof the invention over a DRO is that its resonant structures are notsubstantially susceptible to variations in temperature within a giventemperature range.

The RF/Microwave oscillator may also include a dc biasing circuit forbiasing the active device and preventing RF leakage therethrough. The dcbiasing circuit may also be configured to be frequency adjustable aspreviously discussed. The oscillator may also include a FET gate returncircuit for removing unwanted positive charges being passed through theSchottky junction of the FET during its operation. In addition, theRF/Microwave oscillator may also include a frequency tuning circuit thatis responsive to an input stimuli, such as a tuning voltage, forcontrolling the operating frequency of the oscillator.

BRIEF DESCRIPTION OF THE DRAWINGS

The above mentioned objects, other objects and features of the inventionand the mainer of attaining them will become apparent, and the inventionitself will be best understood by reference to the following descriptionof the preferred embodiments of the invention taken in conjunction withthe accompanying drawings, wherein:

FIG. 1 is a schematic and block diagram of a prior art dielectricresonator oscillator (DRO);

FIG. 2 is a block diagram of a prior art DRO with atemperature-compensating circuit;

FIG. 3 illustrates a cross-sectional view of a dielectric resonatorapparatus using a prior art tuning device;

FIG. 4 is a schematic and block diagram of a DRO in accordance with anaspect of the invention;

FIG. 5 is a schematic and block diagram of an RF amplifier in accordancewith another aspect of the invention;

FIG. 6 is a layout of a direct current (DC) biasing circuit for an RFcircuit in accordance with yet another aspect of the invention;

FIG. 7 is a layout of a DC grounding circuit for an RF circuit inaccordance with another aspect of the invention;

FIG. 8 is a schematic and block diagram of another DRO in accordancewith another aspect of the invention;

FIG. 9 is a layout of a field effect transistor (FET) gate returncircuit in accordance with another aspect of the invention;

FIGS. 10A and 10B are side and bottom cross-sectional views of a cavitydesign for a dielectric resonator in accordance with another aspect ofthe invention;

FIG. 11 is a top view of a dielectric resonator band pass filter inaccordance with another aspect of the invention; and

FIG. 12 is a top view of another dielectric resonator band pass filterin accordance with another aspect of the invention;

FIG. 13 illustrates a cross-sectional view of a dielectric resonatorapparatus using an example tuning device as per an aspect of theinvention;

FIG. 14 illustrates a blow-up view of the tuning portion of the exampletuning device shown in FIG. 13;

FIG. 15 illustrates a cross-sectional view of a dielectric resonatorapparatus using another example of a tuning device as per another aspectof the invention; and

FIG. 16 illustrates a schematic of an example RF/Microwave oscillator inaccordance with another aspect of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Dielectric Resonator Oscillator Using Frequency-Adjustable BiasingCircuits

Referring initially to FIG. 4, a schematic and block diagram of adielectric resonator oscillator (DRO) 100 as an example of a preferredembodiment of the invention is shown. As shown, the DRO 100 used in thisexample of the invention is a series feedback or reflective type of DRO.Although this type of DRO is used to exemplify the invention, it shallbe understood that other types of DROs may be configured to encompassthe invention, including common source, common gate and common drainconfigurations if a field effect transistor is used as the DRO's activedevice; series and parallel dielectric coupling types of DROs; and aparallel feedback transistor DRO. Such types of DRO configurations areshown in the book authored by Darko Kajfex et al., entitled “DielectricResonators,” Artech House 1986, chapter 10.

The DRO 100 preferably includes a metal-semiconductor field effecttransistor (MESFET) or a pseudo-morphic high electron mobilitytransistor (PHEMPT) 102 as the DRO's active device. For the purpose ofexplaining the invention, the active device will be referred to simplyas a field effect transistor or FET 102, and it shall be understood thatit includes at least the above mentioned devices. In addition, it shallbe understood that other active devices may be employed in place of thepreferred MESFET OR PHEMPT, including for example bipolar transistorsand hetero-junction bipolar transistors.

The DRO 100 further includes output and source impedance matchingcircuits 104 and 106 coupled to the drain and source of the FET 102,respectively. The DRO 100 also includes an input resonator transmissionline 108 including an end coupled to the gate of the FET 102. Adielectric resonator puck (DR) 110 is situated a distance from the gateof the FET 102 and spaced apart from the input resonator line 108, as itis conventionally done. In addition, a shunted FET gate return resistorR4 is preferably coupled to opened end of the input resonatortransmission line 108.

As it was previously discussed, there is a need for a DRO design thatcan be easily modified to optimally perform at a plurality of differentdiscreet frequencies within a specified frequency range. The inventionachieves this objective by providing a DRO 100 that it includesadjustable-frequency direct current (DC) biasing circuits, such as drainand source DC biasing circuits 112 and 114. More specifically, the DCbiasing circuits can be easily modified to optimally or substantiallyblock an RF energy or signal cycling at a frequency that comes within atunable range of the biasing circuits. By tuning theadjustable-frequency DC biasing circuits 112 and 114, the DRO 100 can beeasily modified to optimally generate an RF energy or signal cycling atthe specified frequency within a tunable range of the DC biasingcircuits.

The DC biasing circuits 112 and 114 include respective transmissionlines 116 and 118, each preferably having an electrical length of about90 degrees or about an odd multiple thereof (such as 270, 450, 630 . . .etc. degrees) for an RF energy or signal cycling at a frequencypreferably at the upper end of the frequency range for the DRO 100. Inother words, if the DRO 100 can be easily modified to produce outputshaving discreet frequencies within a frequency range from f₁ to f₂, thenit is preferred that the transmission lines 116 and 118 have anelectrical length of about 90 degrees or about an odd multiple thereoffor an RF energy cycling at the upper end of the frequency range, i.e.f₂. In addition, the transmission lines 116 and 118 preferably include ahigh characteristic impedance of, for example, at least about 80 Ohms.

The transmission lines 116 and 118 of the DC biasing circuits 112 and114 each include a first end (RF end) for attachment or electricalconnection with an RF-carrying portion of the DRO 100, preferably nearthe drain and source of the FET 102, respectively. The DC biasingcircuits 112 and 114 further include RF bypass capacitors C3 and C4electrically connected to opposite ends (DC ends) of the transmissionlines 116 and 118. In the preferred embodiment, the capacitance ofcapacitors C3 and C4 should preferably provide a low impedance to groundfor an RF energy cycling at the lower end of the workable frequencyrange, i.e. frequency f₁. In the preferred embodiment, the capacitors C3and C4 are chosen to provide a low impedance having an imaginarycomponent of about one ohm or less at the lower end frequency f₁. The DCbiasing circuits 112 and 114 may also include current-limiting resistorsR5 and R6, which may be variable for properly setting the bias voltagesfor the FET 102.

The DC biasing circuits 112 and 114 become frequency-adjustable by theinclusion of a tuning element, such as tunable transmission lines orstubs 120 and 122. These transmission lines 120 and 122 are tunablebecause their electrical lengths or physical lengths can be adjusted.The tunable transmission lines 120 and 122 include respective first endsattached to about the RF ends of corresponding transmission lines 116and 118 of the DC biasing circuits 112 and 114, respectively. The otherends of the tunable transmission lines 120 and 122 are preferably openedends.

In operation, if the electrical length or physical length of the tunabletransmission lines 120 and 122 is 0 degrees (as if there existed notunable transmission lines), the normalized impedance at the RF ends ofthe transmission lines 116 and 118 are maximized or substantiallymaximized for an RF energy cycling at the upper end frequency f₂. Thisis because without the tunable transmission lines 120 and 122, the DCbiasing circuits 112 and 114 look like the typical quarter wavelengthbiasing lines that are commonly employed in the prior art.

Since transmission lines 116 and 118 can be designed to have a pluralityof characteristic impedances, the normalized impedance parameter is usedbecause it is independent of the characteristic impedance of thetransmission lines, and can be defined as the actual impedance Z dividedby the characteristic impedance of the line Z₀, i.e. Z/Z₀.Alternatively, the maximum normalized impedance can also be defined asan impedance where the phase of an incident RF energy is substantiallyequal and opposite with the phase of a reflected RF energy at the regionpresenting maximum normalized impedance. Such regions in the DRO 100 ofthe invention are the RF ends of the transmission lines 116 and 118, orthe regions where these lines attach to the RF-carrying portion of theDRO 100.

If the electrical lengths of the tunable transmission lines 120 and 122are increased from 0 degrees to above 0 degrees, the maximum normalizedimpedance at the RF ends of the transmission lines 116 and 118 shift forRF energies or signals having frequencies lower than the upper endfrequency f₂. Thus, by increasing or varying the electrical or physicallengths of the tunable transmission lines 120 and 122, the DC biasingcircuits 112 and 114 can be tuned to increase the impedance, preferablyto a substantially maximized normalized impedance, at the RF ends for anRF energy or signal having a discreet frequency within the working rangeof the DRO 100, i.e. frequency range f₁ to f₂.

It is desired for a DRO to have DC biasing circuits that minimizes theeffects they have on the RF energy or signal being produced by the DRO.In order to do this, the DC biasing circuits preferably need to bedesigned to present to the RF circuit or DRO, a substantially maximizednormalized impedance. In this manner, RF energy in the DRO cannot escapethrough or adversely reflect from the DC biasing circuits. For DROs, anyadverse effects from the DC biasing circuits generally increases thephase noise of the RF energy or signal produced by the DRO. Therefore,for performance purposes, the DC biasing circuits 112 and 114 should betuned so that a substantially maximized normalized impedance is presentat the RF ends of the transmission lines 116 and 118 for the desireddiscreet output frequency. Thus, with the use of theadjustable-frequency DC biasing circuits 112 and 114, DRO 100 can beeasily modified to optimally perform at a plurality of discreetfrequencies.

An advantage of the frequency-adjustable DC biasing circuits 112 and 114is that it allows a single DRO design to cover a plurality of discreetfrequencies within the working frequency range. As a result, DROmanufacturers need not custom design the DROs for a particular outputfrequency. This provides for substantial savings in cost, engineeringand manufacturing time, a reduction in inventory and logistics, and alsoimproves the reliability of the DROs.

RF Amplifier and Other RF Devices Using Frequency-Adjustable BiasingCircuits

Although it has been shown that the frequency-adjustable DC biasingcircuits 112 and 114 are substantially beneficial to DROs, it shall beunderstood that these biasing circuits can be employed in other RFcircuits where DC biasing is required. For instance, in FIG. 5, an RFamplifier 150 is shown that includes a plurality of frequency-adjustableDC biasing circuits. As it is conventionally known, the RF amplifier 150preferably includes a MESFET or PHEMPT 152, or can include in placethereof other suitable active devices such as a BJT or a hetero-junctionBJT. The RF amplifier 150 also includes input and output impedancematching circuits 154 and 156, as it is conventionally employed.

The RF amplifier 150 also includes frequency-adjustable drain and sourceDC biasing circuits 157 and 158 and a frequency-adjustable gate DCgrounding circuit 160. As previously discussed, the biasing andgrounding circuits 157, 158 and 160 include respective transmissionlines 162, 164 and 166, each having an electrical length of about 90degrees or about an odd multiple thereof at preferably the upper endfrequency f₄ of the operating frequency range f₃ to f₄ of the amplifier.The transmission lines 162, 164 and 166 include ends (RF ends) forelectrically connecting to the source, drain and gate of the FET 152.

The adjustable-frequency biasing and grounding circuits 157, 158 and 160also include tunable transmission lines or stubs 168, 170 and 172coupled to about the RF ends of the transmission lines 162, 164 and 166.RF bypass capacitors C5 and C6 and current-limiting resistors R7 and R8are coupled to about the DC ends of transmission lines 162 and 164. Adirect path to ground is connected at the DC end of transmission line166 of the grounding circuit 160. By adjusting the lengths of thetunable transmission lines 168, 170 and 172, the biasing circuits 157,158 and 160 can be tuned for any specified frequency within thefrequency range f₃-f₄, such as the center frequency.

Other RF devices or circuits that may benefit from suchfrequency-adjustable biasing or grounding circuits include, for example,mixers, pin attenuators, frequency multipliers or generally any other RFdevice that requires a biasing or grounding circuit where optimum RFisolation at a particular frequency is desired.

Preferred Layout of the Frequency-Adjustable Biasing and GroundingCircuits

Referring to FIG. 6, a preferred configuration for afrequency-adjustable DC biasing circuit 200 is shown. The DC biasingcircuit 200 is preferably formed on a substrate 202, preferably ofalumina material having a preferred height between about 5 to 25 mils,and preferably in a microstrip configuration. The DC biasing circuit 200includes an RF isolating transmission line 204 having an electricallength that extends from an RF end to a DC end of about 90 degrees orabout an odd multiple thereof for an RF energy or signal cycling at apre-determined higher frequency. In the preferred embodiment, the DCbiasing transmission line 204 has a characteristic impedance of at leastabout 80 Ohms, and has a width of about 1 to 2 mils. Also in thepreferred embodiment, the transmission line 204 is formed of two layersof thin films comprised of a lower layer of titanium-tungsten materialand an upper layer of gold material. It shall be understood that thetransmission line 204 can be formed of other suitableelectrically-conductive materials or even of thick film materials.

The DC biasing circuit 200 also preferably includes a thin-filmRF-bypass capacitor 206 comprised of a lower metal layer pad 208including a metallized via hole 209 through the substrate 202 forelectrically connecting to a grounded metal layer (not shown) formed onthe underside of the substrate. The capacitor 206 also includes aninsulating or dielectric layer 210 formed on the metallized pad 208 andmay be comprised of silicon-nitride, silicon-dioxide material, or othersuitable dielectric materials. The capacitor 206 further includes a topmetallization layer 212 being electrically connected to the DC end oftransmission line 204 and also to a biasing or current-limiting thinfilm resistor 214. Preferably, the resistor 214 is formed oftantalum-nitride or nickel chromium, but other suitable materials can beused in place thereof. The thin-film resistor 214 is also connected to aconductive line 216 for receiving a bias voltage V_(B).

The DC biasing circuit 200 also includes at about the RF end of thetransmission line 204, an adjustable-length, open ended transmissionline 218 preferably comprised of a plurality of metallized pads 220arranged along a line or in an array, and preferably electricallyconnected in series by a wire bond or ribbon 222 to form the requiredlength of the open ended transmission line 218. The length of the openended transmission line 218 can be adjusted by adding or deletingmetallized pads to the series of pads electrically connected to eachother by the wire or ribbon bond 222. As it is well known in the art,such wire or ribbon bond are typically welded on such metallized pads220.

Referring to FIG. 7, a preferred configuration for afrequency-adjustable grounding circuit 250 is shown. The groundingcircuit 250 is similar to that of DC biasing circuit 200 described inFIG. 6, in that it includes a metallized transmission line 254preferably formed on an alumina substrate 252 in a microstripconfiguration, and having an electrical length of 90 degrees or an oddmultiple thereof that extends from an RF end to a DC end. The groundingcircuit 250 similarly includes an open ended transmission line 256preferably comprised of a plurality of metallized pads 258, arranged ina line or in an array, and electrically connected in series by wire bondor ribbon 260. The length of the open ended transmission line 256 can beadjusted by adding or deleting metallized pads to the series of padselectrically connected by the wire or ribbon bond 258 for tuning thegrounding circuit for better performance at other RF frequencies.

The grounding circuit 250 differs from the biasing circuit 200 in that adirect path to ground is connected to the DC end of the transmissionline 254, rather than an RF bypass capacitor. Both structures, however,provide a low impedance to ground. In the preferred embodiment, thedirect path to ground comprises a metallized pad 262 connected to aboutthe DC end of the transmission line 254. The metallized pad 262 isgrounded by way of a metallized via hole 264 electrically connected to agrounded metallized layer (not shown) on the underside of the substrate252.

Improved FET Gate Return

Referring now to FIG. 8, a schematic and block diagram of an oscillator,preferably a DRO 300, as an example of another aspect of the inventionis shown. This DRO 300 is directed at improving the FET gate return ofthe prior art to better reduce the phase noise of the DRO. The DRO 300improves the phase noise performance of the device by eliminating thelow resistance FET gate return typically employed in the prior art, andpreferably including a much higher resistance FET gate return. The DRO300 better RF isolates the FET gate resistor by employing a high-Qquarterwave transmission line circuit. The much higher resistancecoupled with the RF isolating circuit reduces the effects the FET gatereturn has on the RF energy or signal of the DRO, thereby reducing thephase noise of the device.

The example DRO 300 of the invention preferably includes a MESFET orPHEMPT 302 as its active device, output and source impedance matchingcircuits 304 and 306 coupled to the drain and source of the FET 302respectively, an input resonator transmission line 308 coupled to thegate of the FET 302, a dielectric resonator puck 309 spaced apart fromand electromagnetically coupled to the input resonator transmission line308, and source and drain bias circuits 310 and 311, which could befrequency-adjustable as previously described or fixed-frequency as inthe prior art biasing circuits.

The improved phase noise performance of the DRO 300 of the inventioncomes about by an improved FET gate return circuit 312. The FET gatereturn circuit 312 comprises a transmission line 314 having an end (RFend) coupled to the input resonator transmission line 308 preferablynear the gate of the FET 302. The transmission line 314 is preferably ofhigh characteristic impedance, for example, of at least about 80 Ohmsand includes an electrical length of about 90 degrees (quarterwavelength) or about an odd multiple thereof (such as 270, 450, 630 . .. etc. degrees) for the operating RF frequency of the DRO 300.

At the opposite end (DC end) of the transmission line 314 is connectedan RF bypass capacitor C9 and a gate return resistor RIO. The bypasscapacitor C9 provides a low impedance to ground at about the DC end ofthe transmission line 314 for an RF energy cycling at the operatingfrequency of the DRO 300. In the preferred embodiment, the impedance ofthe bypass capacitor C9 at the operating frequency of the DRO 300 ispreferably less than about one to two Ohms. The gate return resistor R10preferably includes a resistance of at least about 10 kilo Ohms. Suchhigh value resistance coupled with the RF isolating properties of thequarterwave transmission line 314, provides for minimal effects the gatereturn resistance has on the RF energy or signal of the DRO 300. Thisleads to improved phase noise characteristics of the DRO 300.

The improved FET gate return 312 can also be made to befrequency-adjustable, similar to the adjustable biasing or groundingcircuits previously discussed. In this regard, the transmission line 314preferably includes an electrical length of about 90 degrees (quarterwavelength) or about an odd multiple thereof at the highest operating RFfrequency of the DRO 300, i.e. f₆. In addition, the bypass capacitor C9preferably includes an impedance of less than about one to two Ohms foran RF energy cycling at the lower end of the frequency range, i.e. f₅.In order to optimize the RF isolating properties of the transmissionline 314 for other discreet frequencies, an adjustable-length,preferably open-ended, transmission line 316 is coupled to thetransmission line 314 at about its RF end. As previously discussed, byadjusting the length of the transmission line 316, the RF isolation atthe RF end of the FET gate return 312 can be optimized for otherdiscreet frequencies within the working frequency range f₅ to f₆.

Referring to FIG. 9, a preferred configuration for afrequency-adjustable FET gate return circuit 350 is shown. The FET gatereturn 350 is formed on a substrate 352, preferably of an aluminamaterial having a height of about 5 to 25 mils, and in a microstripconfiguration. The FET gate return 350 includes a transmission line 354having an electrical length from an RF end to a DC end of the line ofabout 90 degrees or about an odd multiple thereof, preferably at theupper frequency of the working frequency range of the DRO. Thetransmission line 354 preferably includes a high characteristicimpedance of at least about 80 Ohms, and therefore has a width of about1 to 2 mils. The transmission line 354 is preferably formed of suitableconducting thin-film material, such as titanium-tungsten-gold orchromium-copper-gold.

The FET gate return 350 further includes an RF bypass capacitor 356coupled to the DC end of the transmission line 354. The capacitor 356comprises a lower metallization layer pad 358 formed on the substrate352, an insulating layer 360, preferably of silicon nitride or silicondioxide material, formed on the lower metallization layer pad, and anupper metallization layer 362 formed on the insulating layer. The lowermetallization layer pad 358 is connected to ground preferably by way ofa metallized via hole 364 formed through the substrate 352 andelectrically connected to a metallization layer (not shown) formed onthe underside of the substrate. The upper metallization layer 362 of thecapacitor 356 is connected to the DC end of the transmission line 354.

The FET gate return 350 further includes a gate return resistor 366electrically coupled to the capacitor 356 by way of the uppermetallization layer 362. The resistor 366 preferably includes aresistance of about at least 10 kilo Ohms and is preferably formed ofsuitable thin-film material, such as tantalum-nitride or nickelchromium. The resistor 366 is electrically connected to ground by way ofmetallization pad 368 including a metallized via hole 370 through thesubstrate 352. The metallized via hole 370 is electrically connected toa grounded metallization layer (not shown) formed on the underside ofthe substrate 352.

In order for the FET gate return 350 to provide optimized RF isolationfor RF frequencies within the working frequency range, anadjustable-length, preferably open-ended transmission line 372 iscoupled to the transmission line 354 at about its RF end. Theadjustable-length transmission line 372 is preferably formed of aplurality of metallized pads 374 formed in a line or in an arrayconfiguration, which are electrically connected to each other by wirebond or ribbon 376. The length of the transmission line 372 can beadjusted by adding or removing metallized pads 374 from the connectionformed by the wire bond or ribbon 376.

Multiple-Frequency Dielectric Resonator Cavity

As previously discussed, it is desired that a DRO be designed such thatit can be easily modified to optimally perform at a plurality ofdiscreet frequencies covering a wide frequency range, preferably byreplacing the appropriate dielectric resonator puck and performingminimal or minor tuning on the DRO circuit. If this can be done, a DROdesign need not be customized for different output frequencies. Thiswould reduce engineering and manufacturing time, inventory, thecomplexity in logistics, and increases the reliability of the DROcircuit.

Not only is it desirable that the DRO be designed so that it is capableof operating over a wide frequency range with minimal modification, itis also desirable that the DRO be temperature stable over such widefrequency range. A dielectric resonator is affected by changes in theenvironment temperature in at least a couple of manners. First, changesin temperature causes the dielectric resonator to either expand orcontract. Such expansion and contraction of a dielectric resonatordecreases and increases its resonant frequency, respectively. Second,the resonant frequency of a dielectric resonator also changes as a metalobject is brought near the dielectric resonator. Specifically, as ametal object is brought near the dielectric resonator, the resonantfrequency of the dielectric resonator typically increases. If it isgradually removed from near the dielectric resonator, the resonantfrequency typically decreases until it is no longer affected by themetal object.

A metal cavity or enclosure that houses a dielectric resonator actssimilar to a metal object in proximity of a dielectric resonator, i.e.it affects the resonant frequency of the dielectric resonator dependingon the distance between the cavity and the dielectric resonator. Asenvironment temperature changes occur, the dielectric resonator cavityor enclosure either expands or contracts. This expansion or contractionof the cavity affects the resonant frequency of the dielectricresonator. Consequently, the resonant frequency of the DRO output issimilarly affected.

In order to improve the temperature stability of a DRO design and alsoprovide a design that is capable of operating over a wide frequencyrange with minimal modifications required, the inventor has conceived anoptimum sizing for a dielectric resonator cavity or enclosure. Thisdiscovered sizing for the dielectric cavity would not only providebetter temperature stability for the DRO, but would achieve suchimproved temperature stability for a plurality of dielectric resonatorshaving resonant frequencies covering a wide frequency range.

Referring to FIGS. 10A and 10B, side and top cross-sectional views of adielectric resonator cavity or enclosure 400 are shown. In the examplecavity of the invention, a cylindrical shaped cavity 400 a and a squareshaped cavity 400 b (preferably square) are used to illustrate theoptimum sizing of the cavity. Reference number 400 is used to refer tothe cavity generally, without regard to its specific shape. The letters“a” and “b” in connection with a reference number are used to refer tothe cylindrical and square cavities, respectively. Although thecylindrical and square cavities are used to exemplify the invention, itshall be understood that the optimum sizing of the cavity can be appliedto other shapes, as explained later.

In detail, the cavity 400 includes an inner surface 402 that ispreferably in a cylindrical 402 a or square 402 b configurationincluding a top or ceiling 404, a bottom or floor 406, and a side wall(cylindrical) or walls (square) 408 and 410. The diameter or width ofthe inner surface 402 of the cavity 400, i.e. the width or diameter ofthe top 404 and bottom 406, can be represented as Dc. Whereas the heightof the inner surface 402 of the cavity 400, i.e. the height of the sidewall(s) 408 and 410, can be represented as Lc.

The dielectric resonator cavity 400 houses or encloses a dielectricresonator structure, preferably of a puck or cylindrical shape. A pairof dielectric resonator pucks 412 and 414 along with correspondingstand-offs 416 and 418 are shown in FIGS. 10A and 10B (FIG. 10B does notshow the stand-offs for the sake of clarity). The dielectric resonatorpucks 412 and 414 represent the preferred extremes of the range ofdielectric resonator size that can be enclosed by the cavity 400 andachieve optimum temperature stability. The width or diameter of adielectric resonator puck that can be housed by the cavity 400 inaccordance with the invention can be represented as Dr. Whereas theheight of such dielectric resonator puck can be represented as Lr.

The inventor has discovered that in order for the cylindrical cavity 400a to provide improved temperature stability for an enclosed dielectricresonator, it is preferred that the diameter Dc of the inner surface 402a of the cavity 400 a, i.e. the width or diameter of the top 404 andbottom 406, be within a range of about 3 times to about 7.5 times thediameter or width Dr of the enclosed dielectric resonator. In addition,it is preferred that the height Lc of the inner surface 402 a of thecavity 400 a, i.e. the height of the side wall 408, be within a range ofabout 3 times to about 7.5 times the height Lr of the encloseddielectric resonator.

For the square cavity 400 b to provide improved temperature stabilityfor an enclosed dielectric resonator, it is preferred that the width Dcof the inner surface 402 b of the cavity 400 b, i.e. the width of thetop 404 and bottom 406, be within a range of about 3 times to about7.5/{square root over (2)} times the diameter or width Dr of theenclosed dielectric resonator. In addition, it is preferred that theheight Lc of the inner surface 402 b of the cavity 400 b, i.e. theheight of the side walls 408 and 410, be within a range of about 3 timesto about 7.5/{square root over (2)} times the height Lr of the encloseddielectric resonator.

In other words, the following preferred relationships hold:

Cylindrical Cavity Square Cavity 7.5 * Dr ≧ Dc ≧ 3 * Dr 7.5/ {squareroot over (2)} * Dr ≧ Dc ≧ 3 * Dr Eq. 1 7.5 * Lr ≧ Lc ≧ 3 * Lr 7.5/{square root over (2)} * Lr ≧ Lc ≧ 3 * Lr Eq. 2

In the example shown in FIGS. 10A and 10B, dielectric resonator puck 412represents the largest dielectric resonator puck that can be preferablyenclosed by cavity 400 and exhibit improved temperature stability. Thatis, the diameter Dc and height Lc of the inner surface 402 of the cavity400 is about 3 (cylindrical and square) times larger than diameter Drand height Lr of the dielectric resonator puck 412. With respect to thelargest dielectric resonator puck 412, it is preferred that the heightof the stand-off 416 is such that the puck 412 is situated near thecenter of the cavity 400. This would result in the distances between top404 and the puck, and the bottom 406 and the puck to be approximatelyequal distances, i.e. k.

The dielectric resonator puck 414 represents the smallest dielectricresonator puck that can be preferably enclosed by cavity 400 and exhibitimproved temperature stability. That is, the diameter Dc and height Lcof the inner surface 402 of the cavity 400 is about 7.5 (cylindrical) or7.5/{square root over (2)} (square) times larger than the diameter Drand height Lr of the dielectric resonator puck 414.

Thus, the preferred frequency range for cavity 400 is determined byabout the resonant frequency of the preferred largest dielectricresonator puck 412 and the preferred smallest dielectric resonator puck414. The resonant frequency for a cylindrical dielectric resonator(puck) may be approximated by the following equation:

f _(RES) =K _(CORR) /[K _(P) ^(½)*(^(π)/₄)^(⅓)*(Dr ² *Lr)^(⅓)]  Eq. 3

where f_(RES) is the resonant frequency of the puck, K_(CORR) is aconstant correction factor, and K_(P) is the dielectric constant of thepuck material. Most cylindrical dielectric resonators are designed suchthat their height Lr fall within the following range relative to theirdiameters Dr:

0.35*Dr<Lr<0.45*Dr  Eq. 4

Assuming that the height of a dielectric resonator puck is about 0.40times its diameter Dr, equation 3 reduces to the following relationship:

f _(RES) =K _(CORR) /[K _(P) ^(½)*(^(π)/₄)^(⅓)*(0.4)^(½) *Dr]  Eq. 5

Equation 5 illustrates that there exists a linear, but inverse,relationship between the diameter Dr of the dielectric resonator puckand its resonant frequency f_(RES). Assuming that the largest andsmallest dielectric resonator pucks 412 and 414 have the same dielectricconstant, then the working frequency range f₇ to f₈ of the cavity 400can be approximated by the following relationship:

f ₈ /f ₇ =f _(RES)(small puck)/f _(RES)(large puck)=  Eq. 6

Dr (large puck)/ Dr(small puck)

Substituting the preferred ranges for the diameter Dr of encloseddielectric resonator pucks given by equation 1, the followingrelationship approximates the working frequency range for the cavity400:

Cylindrical Cavity Square Cavity f₈ / f₇ = 7.5/3 = 2.5 f₈ / f₇ =7.5/3{square root over (2)} = 1.77 Eq. 7

Hence, according to equation 7, it can be seen that the workingfrequency range for the cylindrical cavity 400 a is well over an octave,and for the square cavity is about three-quarters of an octave. As aresult, a plurality of DROs having different respective discreet outputfrequencies covering a wide frequency range can be designed with the useof a single cavity 400. This is of considerable advantage with respectto engineering and manufacturing time, cost, inventory, logistics andreliability.

Although the preferred dimensions for the cavity 400 is provided byequations 1 and 2, it shall be understood that other preferred rangesmay include any subset within the preferred range. For instance, thefollowing Tables 1 and 2 provide examples of other preferred rangeswithin those given by equations 1 and 2 for the cylindrical and squarecavities, respectively:

TABLE 1 (Cylindrical Cavity) Dc Dc Lc Lc (Approximate (Approximate(Approximate (Approximate Lower Value) Upper Value) Lower Value) UpperValue) 4.5 * Dr 7.5 * Dr 4.5 * Lr 7.5 * Lr 5.0 * Dr 7.5 * Dr 5.0 * Lr7.5 * Lr 5.5 * Dr 7.5 * Dr 5.5 * Lr 7.5 * Lr 6.0 * Dr 7.5 * Dr 6.0 * Lr7.5 * Lr 5.0 * Dr 7.0 * Dr 5.0 * Lr 7.0 * Lr 5.0 * Dr 6.5 * Dr 5.0 * Lr6.5 * Lr

TABLE 2 (Square Cavity) Dc Dc Lc Lc (Approximate (Approximate(Approximate (Approximate Lower Value) Upper Value) Lower Value) UpperValue) 3.50 * Dr 5.30 * Dr 3.50 * Lr 5.30 * Lr 4.00 * Dr 5.30 * Dr4.00 * Lr 5.30 * Lr 4.50 * Dr 5.30 * Dr 4.50 * Lr 5.30 * Lr 5.00 * Dr5.30 * Dr 5.00 * Lr 5.30 * Lr 4.00 * Dr 4.80 * Dr 4.00 * Lr 4.80 * Lr4.00 * Dr 4.30 * Dr 4.00 * Lr 4.30 * Lr

As previously mentioned, the optimum sizing for the dielectric resonatorcavity can be applied to other shapes. As a general rule, the distancefrom the center of the shape (where the dielectric resonator lies) tothe most outer edges of its inner surface should not exceed thecorresponding distance of the upper boundary for the cylindrical cavity,i.e. 7.5/2*Dr. Also, that same distance should not be less than thecorresponding distance of the lower boundary for the cylindrical cases,i.e. 3/2*Dr.

Although the dielectric resonator cavity 400 has been described for DROapplications, it shall be understood that it can also be used for otherdielectric resonator applications, such as filters. Referring to FIG.11, a top view of an example of a dielectric resonator band pass filter450 in accordance with the invention is shown. For illustrative purposesonly, the filter 450 is made to be a band pass filter having four poles.It shall be understood, however, that the number of poles, the filtertype (e.g. band pass), or configuration of the filter is not critical tothe invention. Thus, it can encompass band reject filters or other typesof dielectric resonator filters.

The filter 450 includes a metallized or electrically conducting housing451. Being a four-pole filter, the filter 450 includes four dielectricresonators 452 a-d along with corresponding stand-offs 454 a-454 d(shown as dashed lines because they underlie the resonators) situatedwithin a cavity 455 of the housing 451. The filter 450 is preferablyconfigured such that adjacent dielectric resonators areelectromagnetically coupled to each other.

The dielectric resonator filter 450 may also include an inputtransmission line 456 preferably formed on a substrate 458 in amicrostrip configuration. The substrate 458 may be situated on a raisedportion 459 on the housing 451, and alternatively include a carrier (notshown). The input transmission line 456 may also be coupled to a coaxialconnector 460 that is connected on the housing 451, as it isconventionally done. Preferably, the dielectric resonator 452 a alongwith its stand-off 454 a can be situated on top of the substrate 458such that a portion of the transmission line 456 underlie the dielectricresonator. Also in the preferred embodiment, a region of thetransmission line 456 that is about electrically 90 degrees or an oddmultiple thereof from the end of the line is situated below thedielectric resonator 452 a, since that is the region that achieves theoptimum coupling between the line and the resonator.

The dielectric resonator filter 450 may also include an outputtransmission line 462 preferably formed on a substrate 460 in amicrostrip configuration and situated on raised portion 465 of thehousing 451. These elements can be configured similar to the input ofthe filter including a coaxial connector 468 coupled to the transmissionline 462. In actuality, the input and output may be a misnomer since thefilter is a passive device, and the input can serve as the output andvice-versa.

The middle dielectric resonators 452 b-c including their respectivestand-offs 454 b-c may be situated in a recess portion 470 of thehousing 451 so that preferably all of the dielectric resonators aresituated at the same height. This provides for optimum coupling betweenadjacent resonators. The dielectric resonators 452 a-d may be connectedto the corresponding standoffs 454 a-d and together to the housing 451and/or substrates 458 and 464 by suitable adhesives or mechanicaldevices.

In order to provide improved temperature stability for the dielectricresonator filter over a wide range of frequencies, the cavity 455 of thehousing 451 is preferably dimensioned with respect to the dielectricresonators 452 a-d in accordance with ranges specified by equations 1and 2, or with the subset ranges provided by Tables 1 and 2. This wouldallow one housing design to be used on a multitude of filters havingdifferent frequency responses within the working frequency range of thecavity.

Referring to FIG. 12, a top view of another example of a dielectricresonator filter 500 in accordance with the invention is shown. Thefilter 500 is similar to filter 450, except each of the dielectricresonators are preferably enclosed within respective cavities andcoupling between resonators is preferably by way of microstriptransmission lines, feedthroughs or the like.

More specifically, the filter 500 is also a four-pole filter in that itincludes four dielectric resonators 504 a-d, preferably of cylindricalshape, and preferably mounted on respective standoffs 506 a-d (shown asdotted lines since they preferably underlie the respective pucks). Thedielectric resonators 504 a-d along with respective standoffs 506 a-dare preferably mounted together and to the substrates 510 a-d withsuitable adhesives or mechanical devices. The substrates 510 a-dpreferably include respective input and output transmission lines 508a-d, preferably formed on the substrates as metallized thin-films in amicrostrip configuration. As mentioned above, the fact that thetransmission lines 508 are designated as input and output is a misnomersince this filter is a passive device, and therefore the inputs canserve as outputs, and vice-versa.

The housing 502 of the dielectric resonator filter 500 preferablyincludes four cavities 512 a-d for enclosing each of the respectivedielectric resonators 504 a-d. The cavities 512 a-d are preferablyseparated by traversing metallized walls 514 a-514 c which arepreferably formed integral as part of the housing 502. The dimensions ofthe cavities 512 a-d with respect to the corresponding dielectricresonators 512 a-d are preferably designed in accordance with the rangesprescribed by equations 1 and 2, or any other the sub-ranges or subsetswithin the ranges prescribed by equations 1 and 2, such as those listedin Tables 1 and 2. As previously discussed, such cavity design providesfor improved temperature stability over a wide frequency band, whichallows a single cavity design to be used on a plurality of filtershaving frequency responses within the band.

The traversing walls 514 a-c include respective openings 516 a-c thateach preferably include a feedthrough, transmission line or any couplingdevice for coupling between adjacent dielectric resonators 504. Thefilter 500 may also include suitable input and output connectors, suchas coaxial connectors 520 and 522.

Temperature-Compensating Resonant Frequency Tuning Devices

FIG. 13 shows a cross-sectional view of a dielectric resonator apparatus550 using a dielectric resonator tuning device 560 in accordance with anaspect of the invention. The term dielectric resonator apparatus 550 isused herein to refer to generally a device or circuit that requires adielectric resonator in performing its operation. Such dielectricresonator apparatus include dielectric resonator oscillators DROs anddielectric resonator filters, such as the examples shown in FIGS. 1, 2,4, 8, 11 and 12. However, other apparatus, circuits or devices that usea dielectric resonator can also benefit from the use of the tuningdevice 560 of the invention. Typical of dielectric resonator devices orcircuits, the dielectric resonator apparatus 550 includes a housing 552preferably configured to form an enclosed cavity 554. Within the cavity554 are situated a dielectric resonator 556 and stand-off 558, which arepreferably mounted on the bottom of the housing 552 or on top of asubstrate (not shown).

The example dielectric resonator tuning device 560 of the inventionpreferably comprises several components, including a centrally-locatedtuning element or slug 568, a head portion 562, an inner sleeve 570, andan outer sleeve 572. The tuning element 568 is the component of thetuning device 560 that electromagnetically interacts with the dielectricresonator so that tuning and temperature stability of the resonantfrequency of the dielectric resonator is provided. It is the element ofthe tuning device 560 that moves towards and away from the dielectricresonator 556 to shift its resonant frequency so that tuning andtemperature compensation of the dielectric resonator apparatus 550 isachieved.

In the preferred embodiment, the tuning device or slug 568 is preferablyin the shape of an elongated solid cylinder. However, it shall beunderstood that the shape of the tuning element or slug 568 is notcritical to the invention, and can encompass other shapes, like solidpolygonal such as rectangular, pentagon, hexagon, and so forth. Thetuning element or slug 568 also need not be completely solid. In thepreferred embodiment, threads are formed as part of the side walls ofthe tuning element 568, preferably extending from about its upper end topass its middle region.

In addition to the tuning element 568, the tuning device 560 includesthe head portion 562 that is preferably coaxially attached to the top ofthe tuning element. The head portion 562 allows for receipt of amechanical device for use in positioning the end of the tuning elementwith respect to the dielectric resonator puck 556. In the preferredembodiment, the head portion 562 includes a slot 564 at its top portionthereof which is sized and dimensioned to receive a flat head screw foruse in rotating the head portion and the tuning element 568. It shall beunderstood that the particular mechanical device used in positioning thetuning element 568 is not critical to the invention, and consequently,the head portion 564 can be designed to accept other types of mechanicaldevices, such as Phillips and Hex screw drivers, to name a few. Also inthe preferred embodiment, the head portion 562 may also include threads566 which are preferably configured to thread with the threads of thetuning element 568.

Along with the tuning element 568 and the head portion 562, the tuningdevice 560 includes an inner sleeve 570. As will be explained in moredetail later, the inner sleeve 570 can be used to augment the movementof the tuning element 568 with respect to the dielectric resonator 556with temperature variation. In addition, the inner sleeve 570facilitates the positioning of the end of the tuning element 568 at thedesired distance (“sweet spot”) from the dielectric resonator puck 556.In the preferred embodiment, the inner sleeve 570 is in the shape of ahollow cylinder, including threads formed as part of its inner wall thatare configured to rotatably mate with the threads of the head portion562 and the tuning element 568. Another set of threads are preferablyformed as part of the outer wall of the inner sleeve 570.

In addition to the tuning element 568, head portion 562, and innersleeve 570, the tuning device 560 also includes the outer sleeve 572. Aswith the inner sleeve 570, the outer sleeve 572 can be used to augmentthe movement of the tuning element 568 with respect to the dielectricresonator 556 with temperature variation. In addition, the outer sleeve572 also facilitates the positioning of the end of the tuning element568 at the desired distance (“sweet spot”) from the dielectric resonator556. A further function of the outer sleeve 572 is that it provides fora rotatable attachment of the tuning device 560 to the housing 552.

In the preferred embodiment, the outer sleeve 572 is in the shape of ahollow cylinder, including threads formed as part of its inner wall thatare configured to rotatably mate with the outer threads of the innersleeve 570. Another set of threads are preferably formed as part of theouter wall of the inner sleeve 570 and are configured to rotatablyengage with the housing 552 and a nut 572. The outer sleeve 570additionally includes a lip portion 573 preferably formed at its upperend. The lip portion 573 is configured to engage with a mechanicaldevice or human digits to facilitate rotation of the outer sleeve 572,and also to prevent rotation of the outer sleeve while the head portion562/tuning element 568 and/or the inner sleeve 570 are being rotated.The lip portion 573 also serves as a stop to limit the movement of thetop end of the outer sleeve 572 into the housing 552 and or cavity 554when the lip portion abuts the Hex nut 574, or housing if the nut is notpresent. The nut 574, preferably a Hex nut, facilitates the securedattachment of the tuning device 560 to the housing 552 of the dielectricresonator apparatus 550. A tuning disk (not shown) may also beconcentrically attached to the end of the tuning element 568 in order toprovide a larger surface to interact with the electromagnetic wavesemanating from the dielectric resonator 556.

As one of its functions, the example tuning device 560 of the inventionfacilitates in the tuning or selection of the desired resonant frequencyof the dielectric resonator 556. Typically in the design of a dielectricresonator apparatus, the dielectric resonator is selected in a mannerthat its “stand alone” or unaffected resonant frequency is slightlylower than the desired resonant frequency for the resonator. Theresonant frequency of a dielectric resonator can be increased bybringing a metal object in proximity to the resonator. The tuningelement 568 of the tuning device 560 serves as the metal object forincreasing the resonant frequency of the dielectric resonator 556. Inoperation, the movement of the tuning element 568 towards the dielectricresonator 568 increases the resonant frequency of the resonator.

With the tuning device 560 of the invention, three separate operationscauses the movement of the tuning element 568 with respect to thedielectric resonator. First, rotating the head portion 562/tuningelement 568 while keeping the inner and outer sleeves stationary, causesmovement of the tuning element towards or away from the dielectricresonator 556. Second, rotating the inner sleeve 570 and the headportion 562/tuning element 568 together, while keeping the outer sleeve572 stationary, causes movement of the tuning element towards or awayfrom the dielectric resonator. Third, rotating the head portion562/tuning element 568 together with the inner and outer sleeves570-572, causes movement of the tuning element towards or away from thedielectric resonator 556.

One advantage of having three separate operations for controlling theposition of the tuning element 568 with respect to the dielectricresonator is that the three rotating operations discussed above can bemade to provide coarse, medium and fine movement of the tuning elementwith respect to the dielectric resonator. For example, the threads ofthe head portion 562/tuning element 568 and the inner threads of theinner sleeve 570 can be made to have a relatively small pitch so thatrotation of the head portion/tuning element provide fine movement of thetuning element towards and away from the dielectric resonator. The outerthreads of the inner sleeve 570 and the inner threads of the outersleeve 572 can be made to have a medium-size pitch so that rotation ofthe head portion/tuning element and inner sleeve provide medium movementof the tuning element 568 towards and away from the dielectricresonator. The outer threads of the outer sleeve 572 and the top wall ofthe housing 552 can be made to have a relatively large pitch so thatrotation of all the elements together causes coarse movement of thetuning element 568 towards and away from the dielectric resonator.

Not only does the example tuning device 560 of the invention serve toadjust the position of the end of the tuning element 568 with respect tothe dielectric resonator in order to tune its resonant frequency, it canalso serve to stabilize the resonant frequency of the dielectricresonator with changes in the environment temperature. For example, astemperature increases, the dielectric resonator 556 expands. Since theresonant frequency of a dielectric resonator is inversely proportionalto the size of the dielectric resonator, the expansion of the dielectricresonator 556 causes its resonant frequency to decrease. Thus, astemperature increases, the resonant frequency of the dielectricresonator decreases. Similarly, as temperature decreases, the dielectricresonator 556 contracts, and consequently causes the resonant frequencyto increase.

Also with an increase in temperature, the housing 552 of the dielectricresonator apparatus 550 expands. This causes the movement of the housingwalls away from the dielectric resonator 556. Since bringing a metalobject in proximity to a dielectric resonator tends to increase itsresonant frequency, the housing walls moving away from the dielectricresonator as temperature increases tends to lower the resonantfrequency. Thus, as temperature increases, there are two actions thattend to cause the resonant frequency of the dielectric resonator 556 todecrease, i.e. the expansion of both the dielectric resonator and thehousing 552. Similarly, as temperature decreases, there are two actionsthat tend to cause the resonant frequency of the dielectric resonator toincrease, i.e. the contraction of the dielectric resonator and thecontraction of the housing 552.

To counteract these two actions that causes variation in the resonantfrequency of the dielectric resonator 556 as temperature changes, theexample tuning device 560 of the invention includes three elements thatcan assist in stabilizing the resonant frequency with temperaturechanges. These elements include the tuning element 568, the inner sleeve570, and the outer sleeve 572. In operation, as temperature rises withinthe operating temperature range of the dielectric resonator apparatus,the tuning element 568, inner sleeve 570, and outer sleeve expandlinearly with temperature. The expansion of these elements causesmovement of the end of the tuning element 568 towards the dielectricresonator 556. Because the tuning element 568 acts as a metal object inproximity to a dielectric resonator, the movement of this elementtowards the resonator tends to raise or increase the resonant frequencyof the dielectric resonator 556, which counteracts the actions of theexpansion of the dielectric resonator and housing 556 in tending tolower or decrease the resonant frequency.

Similarly, as temperature decreases within the operating temperaturerange of the dielectric resonator apparatus 550, the tuning element 568,inner sleeve 570, and outer sleeve 572 contract linearly withtemperature. The contraction of these elements causes movement of theend of the tuning element 568 away from the dielectric resonator 556.Because the tuning element 568 acts as a metal object in proximity to adielectric resonator, the movement of this element away from thedielectric resonator tends to lower or decrease the resonant frequencyof the dielectric resonator 556, which counteracts the actions of thecontractions of the dielectric resonator 556 and housing 552 that tendto increase the resonant frequency.

One particular advantage of the example tuning device 560 of theinvention is the large range of movement of the tuning element 568 withchanges in temperature that can be achieved with the device. The reasonfor this is that the cumulative expansions of the outer sleeve 572,inner sleeve 570 and the tuning element 568 can be made to contribute tothe movement of the tuning element 568. Since three elements can be madeto contribute to the movement of the end of the tuning element 568 withrespect to the dielectric resonator 556, a large range of movement canbe achieved. In the preferred embodiment, the selection of the materialsfor the outer sleeve 572, inner sleeve 570, and tuning element 568 issuch that the range of movement of the tuning element achieved isgreater than what is required to provide the desired temperaturestability of the resonant frequency of the dielectric resonator 556. Inother words, the materials are selected to overcompensate for theresonant frequency drift due to temperature changes. Proper adjustmentof the tuning device 560 to provide the desired temperature compensationis explained as follows.

Another advantage of the example tuning device 560 of the invention isits versatility and ease in obtaining the desired movement of the end ofthe tuning element 568 to provide the desired temperature stability ofthe resonant frequency of the dielectric resonator 556. The movement ofthe end of the tuning element 568 is not only governed by thetemperature expansion coefficients of the outer sleeve 572, inner sleeve570, and tuning element 568, it is also governed by the length of theirrespective materials below their respective contacting points to theirrespective adjacent outer elements. FIG. 14 is used to better explainthis principle.

FIG. 14 shows a “blow up” view of the lower portion of the exampletuning device 560 of the invention shown in FIG. 13. The threads of thehousing 552, outer sleeve 572, inner sleeve 570, and tuning element 568are not shown for simplicity sake. In this simplified view of theexample tuning device 560, the regions of attachment or abutment of theelements to adjacent elements can be represented by nodes, i.e.equivalent single contact points. For instance, the attachment orabutment of the outer sleeve 572 with the housing 552 can be representedas node 576. Similarly, the attachment or abutment of the inner sleeve570 with the outer sleeve 572 can be represented by node 578. Likewise,the attachment or abutment of the tuning element 568 with the innersleeve 568 can be represented by node 580.

During temperature changes, the regions of attachment or abutments ofthe various elements of the tuning device 560, i.e. nodes 576, 578 and580, typically remain substantially fixed or stationary with respect toeach other, along the longitudinal axis of the elements. However, forregions of the elements not at the attachment or abutment regions, suchas the regions below the nodes 576, 578, and 580, the regions expand orcontract in the longitudinal direction relative to the nodes. Thisexpansion or contraction of these regions of the elements below thenodes is a function of the temperature expansion coefficients of theirrespective materials, and also of their respective lengths. For example,the outer sleeve 572 expands or contracts with changes in temperaturebelow node 576 substantially linearly proportional to the length L1.Similarly, the inner sleeve 570 expands or contracts with changes intemperature below node 578 substantially linearly proportional to thelength L2. Likewise, the tuning element 568 expands or contracts withtemperature below node 580 substantially linearly proportional to thelength L3.

The desired movement of the tuning element 568 with changes intemperature can be achieved by adjusting the lengths L1-3 of the outersleeve 572, inner sleeve 570, and tuning element 568 that lie belownodes 576, 578 and 580 respectively. This can be easily performed withthe example tuning device 560 of the invention by rotating the outersleeve to change length L1, rotating the inner sleeve 570 with respectto the outer sleeve 572, or vice-versa, to change length L2, androtating the head portion 562/tuning element 568 with respect to theinner sleeve, or vice-versa, to change length L3. Since the tuningdevice 560 provides three variables for controlling the rate of movementof the tuning element 568 with changes in temperature, precision as tothe desired movement can be achieved.

FIG. 15 shows a cross-sectional view of another dielectric resonatorapparatus 600 that uses another embodiment of a tuning device 610 inaccordance with the invention. Similar to the example dielectricresonator apparatus 550 of the invention, the example dielectricresonator apparatus 600 of the invention includes a housing 602preferably configured to form a cavity 604 for housing a dielectricresonator 606 and corresponding stand-off 608 preferably situated at thebottom wall of the housing, or alternatively, on a substrate (notshown). Similar to the example tuning device 560 of the invention, theexample tuning device 610 of the invention includes a head portion 612having a slot 614 and threads 616, a tuning element or slug 618 attachedto or integral with the head portion 616, and inner and outer sleeves620 and 624. A nut 626, preferably a Hex nut, is also provided tosecurely attach the tuning device 610 to the housing 602.

In addition to the inner and outer sleeves 620 and 624, the exampletuning device 610 of the invention further includes an intermediatesleeve 622 situated between the inner and outer sleeves. In thepreferred embodiment, the sleeves 620, 622, and 624 and the tuningelement 618 are substantially cylindrically shaped and concentric witheach other. The tuning element 618 is preferably solid throughout,whereas the sleeves 620, 622, and 624 are configured as hollow shells toencompass the adjacent sleeve(s) and/or tuning element. Also the threadsof the sleeves 620, 622 and 624 and the tuning element 618 can havedifferent pitches to achieve coarse, medium, or fine movement of thetuning device by rotation of these elements as discussed above.

One advantage of the additional intermediate sleeve 622 is that an evenlarger range of movement of the tuning element 618 with changes intemperature can be potentially achieved over that of the tuning device610. The reasons being is that there are four elements in the tuningdevice 610 contributing to the movement of the tuning element 618,namely, the sleeves 620, 622 and 624, and the tuning element 618.Another advantage of the example tuning device 610 of the invention isthat it includes four parameters for adjustment to precisely provide thedesired movement of the tuning element 618 with temperature, so that thedesired temperature stability of the resonant frequency of thedielectric resonator 606 is achieved. These parameters includes thelengths of the sleeves and tuning element below there respective outwardadjacent contact points or nodes. It shall be understood that additionalsleeves may be added to the tuning device 610 in order to provide alarger range of movement of the tuning element and better control on therate of its movement with changes in temperature.

The tuning devices 550 and 610 of the invention can be used on acircuit, device or apparatus that includes a dielectric resonator inperforming its functions. Such circuit, device or apparatus can includedielectric resonator oscillators (DROs), such as the ones describedherein and/or depicted in FIGS. 1, 2, 4 and 8, to name a few. Suchcircuit, device or apparatus can also include dielectric resonatorfilters, such as the ones described herein and/or depicted in FIGS.11-12, to name a few. The tuning devices 550 and 610 can also be usedwith the multiple-frequency dielectric resonator cavity 400 of theinvention in order to be used with a line of dielectric resonatorapparatus operating at a plurality of distinct frequencies or frequencyranges.

High-Q and Low Loss RF/Microwave Oscillator with an Inherent TemperatureStability Property

As discussed earlier, dielectric resonator oscillators (DROs) have grownin popularity because of their superior high-Q, low-loss, and low phasenoise properties. However, their attractiveness is somewhat tarnishedbecause of their inherent susceptibility to environment temperaturevariation. What would be optimal is to have an RF/Microwave oscillatorthat has the high-Q, low-loss, and low phase noise characteristics of aDRO, with the additional characteristic that it is not highlysusceptible to temperature variation, or that it has an inherenttemperature stable property. Such an oscillator would reduce engineeringtime in avoiding design of temperature compensating techniques, reducemanufacturing time in foregoing the implementation of temperaturecompensation devices, circuits or structures, reduce inventory andlogistics due to less components, and increase reliability also due toless components. Such an oscillator is provided herein as follows.

FIG. 16 shows a schematic of an example RF/Microwave oscillator 650 inaccordance with an aspect of the invention. The RF/Microwave oscillator650 preferably includes an active device 652, a drain output linecircuit 654, a series feedback line circuit 656, at least one biascircuit 658 for the active device, a resonator line 660, a tune line orresonator circuit 662, a FET gate return circuit 664, and a frequencytuning circuit 668. In the preferred embodiment, the active device 652comprises a metal-semiconductor field effect transistor (MESFET), butcan include other types of active devices. For example, the activedevice 652 can include a high electron mobility transistor (HEMPT) ormore specifically, pseudo-morphic high electron mobility transistor(PHEMPT). Alternatively, the active device 652 can be of a bi-polartransistor type, such as a hetero-junction bi-polar transistor. Theparticular active device is not critical to the invention.

In the preferred embodiment, the drain (D) of the MESFET 652 is coupledto the drain output coupled line circuit 654. The drain output coupledline circuit 654 is preferably comprised of at least a pair of coupledtransmission lines. However, if improved performance is desired, as willbe explained later, multiple cascaded pairs of coupled transmissionlines can be employed. The example RF/Microwave oscillator 650 of theinvention is shown to have a drain output line circuit 654 thatcomprises two cascaded pairs of coupled transmission lines. The drainoutput line circuit 654 is preferably designed to provide substantiallyoptimal impedance matching and coupling to an output transmission line670 at the operating frequency of the oscillator 650. If the activedevice 652 is not a FET or HEMPT, the appropriate terminal of the deviceis to be coupled to the drain output line circuit 654, as understood bythose skilled in the art.

Also in the preferred embodiment, the source (S) of the MESFET 652 iscoupled to the series feedback line circuit 656. The series feedbackline circuit 656 is preferably comprised of at least a pair of coupledtransmission lines with an open end. However, if improved performance isdesired, as will be explained later, multiple cascaded pairs of coupledtransmission lines can be employed. The example RF/Microwave oscillator650 of the invention is shown to have a series feedback line circuit 656that comprises two cascaded pairs of coupled lines. The feedback seriesline circuit 654 is preferably designed to provide optimal coupling backof RF energy to the input of the oscillator at the operating frequencyof the oscillator 650. If the active device 652 is not a FET or HEMPT,the appropriate terminal of the device is to be coupled to the seriesfeedback line 656, as understood by those skilled in the art.

In order to provide a bias voltage for the MESFET 652 of the exampleRF/Microwave oscillator 650 of the invention, the bias circuit 658 isincluded to supply the needed direct current (dc) bias voltage to theMESFET with minimal effects on the RF/Microwave energy generated by theoscillator. This is typically done by including a quarterwave linehaving one end connected to the source (S) of the MESFET 652 and theother end connected to an RF bypass capacitor coupled to ground viasuitable means, such as a via hole. As it is conventionally known, thebias voltage is applied to the node interconnecting the quarterwave lineand the RF bypass capacitor. In the alternative, the bias circuit 658can be made frequency-adjustable in the same manner asfrequency-adjustable biasing circuits described herein and/or depictedin FIGS. 4-9. Similar bias circuits can also be employed for supplyingdc bias voltage to other terminals of the MESFET, if desired. If theactive device 652 is not a FET or HEMPT, the appropriate terminal of thedevice is to be coupled to the bias circuit 658, as understood by thoseskilled in the art.

In the preferred embodiment, the gate (G) of the MESFET 652 is coupledto the FET gate return circuit 664. As explained earlier, the gatereturn circuit 664 provides a path to ground for unwanted positivecharges that pass through the Schottky diode junction of the gate duringthe operation of the MESFET. The gate return circuit 664 preferablycomprises a quarterwave line coupled to a gate return resistor and an RFbypass capacitor, both of which are coupled to ground. Preferably, thegate return resistor is characterized by a high resistance, such as, forexample, about 10 kilo Ohms. It is also preferred that the RF bypasscapacitor have a susceptance that does not exceed about one to two Ohms.Alternatively, the gate return resistor circuit 664 can befrequency-adjustable similar to the gate return circuits 312 and 350described herein and shown in FIGS. 8 and 9.

In the preferred embodiment, the gate (G) of the MESFET 652 is coupledto the input resonator line 660, as it is conventionally done for DROs.The input resonator line 660, in turn, is preferably coupled to a tuneline or resonator circuit 660 comprising at least a pair of coupledtransmission lines. If improved performance in the form of higher-Q andlow-loss is desired, it is preferred that multiples cascaded pairs ofcoupled transmission lines are used to configure the resonatorstructure. The example RF/Microwave oscillator 650 of the invention isshown to have a tune line of resonator circuit 66 that includes threecascaded pairs of coupled transmission lines. The coupled transmissionlines are preferably designed to resonate at the operating frequency ofthe oscillator 650. If the active device 652 is not a FET or HEMPT, theappropriate terminal of the device is to be coupled to the inputresonator line 660, as understood by those skilled in the art.

In the preferred embodiment, the tune line or resonator circuit 660 iscoupled to the frequency tuning circuit 668. The frequency tuningcircuit 668 allows external control of the resonant frequency of thetune line or resonator circuit 662. Preferably, the stimuli forproviding the external frequency control is a tuning voltage. In thepreferred embodiment, the frequency tuning circuit 668 comprises avaractor diode having an anode coupled to the tune line or resonatorcircuit 662 and a quaterwave line coupled to ground. The cathode of thevaractor is coupled to an RF bypass capacitor connected to ground and toa second quarterwave line. The second quaterwave line is, in turn,coupled to a tune voltage input line and a second RF bypass capacitorconnected to ground. The tuning voltage is applied to the frequencytuning circuit 668 by way of the tune voltage input line. Adjustment ofthe tuning voltage causes changes in the impedance of the varactordiode, which in turn, affects the resonant frequency of the tune line orresonator circuit 662.

In the preferred embodiment, the various circuits of the RF/Microwaveoscillator 650, namely the drain output line circuit 654, the seriesfeedback line circuit 656, the biasing circuit 658, the gate returncircuit 664, the input resonator line 660, the tune line or resonatorcircuit 662, and the frequency tuning circuit 668 may be formed on asuitable substrate, such as alumina, quartz, Duriod® or semiconductorsubstrates, preferably in a microstrip form. In this configuration, thequarterwave, coupled and input resonator transmission lines arepreferably formed as metallized thin-film layers deposited on thesubstrate. The RF capacitors may of a thin-film overlay type,interdigital type, chip type or other suitable types formed or mountedon the substrate. The grounds are preferably formed of metallized viaholes, but can include wrap-arounds, wire or ribbon bond connection to agrounded surface. It shall be understood that these circuits can also beconfigured into other mediums, including stripline, co-planermicrostrip, and suspended stripline, to name a few.

The advantage of the example RF/Microwave oscillator 650 of theinvention is that it can be easily configured to perform with a high-Q,low-loss, and low phase noise characteristic comparable to that of aDRO. Specifically, the overall Q of the RF/Microwave oscillator 650 isaffected by the individual Qs of the tune line or resonator circuit 662,drain output line circuit 654, and the series feedback line circuit 656.The Qs of these circuits are, in turn, proportional to the number ofcascaded pairs of coupled transmission lines. That is, the more cascadedpairs of coupled transmission lines in these circuits, the higher the Qfor each of these circuits. By including a sufficient number of cascadedpairs of coupled transmission lines for each of these circuits, thehigh-Q, low-loss and phase noise performance of a DRO can be achieved.

Such as RF/Microwave oscillator 650 enjoys the performancecharacteristic of a DRO, without the adverse affect of having adielectric resonator that has a resonant frequency highly susceptible totemperature variations. Since a dielectric resonator is not required bythe RF/Microwave oscillator 650, there is no need to provide temperaturecompensation devices or structure to achieve the desired temperaturestability. This translates to reduce engineering time, manufacturingtime, cost, inventory, logistics and an increase in the reliability ofthe oscillator.

While the invention has been described in connection with variousembodiments, it will be understood that the invention is capable offurther modifications. This application is intended to cover anyvariations, uses or adaptation of the invention following, in general,the principles of the invention, and including such departures from thepresent disclosure as come within known and customary practice withinthe art to which the invention pertains.

It is claimed:
 1. A temperature-compensating dielectric resonator tuningdevice comprising: a tuning element including a first portion thereoffor electromagnetically interacting with a dielectric resonator, whereinsaid tuning element is an elongated shaft and said first portion is saidend of said shaft; a first material disposed on said tuning element forcausing said portion of said tuning element to move with respect to saiddielectric resonator as temperature varies in accordance with a firstlinear temperature expansion coefficient, wherein said first material isconfigured into a first sleeve disposed coaxially around said shaft; anda second material disposed on said first material for causing saidportion of said tuning element to move with respect to said dielectricresonator as temperature varies in accordance with a second lineartemperature expansion coefficient, wherein said second material isconfigured into a second sleeve disposed coaxially around said firstsleeve; wherein said shaft includes a first set of threads formed aroundsaid shaft and configured to mate with a second set of threads formed onan inner portion of said first sleeve, and wherein said first sleeveincludes a third set of threads formed on an outer portion thereof, andconfigured to mate with a fourth set of threads formed on an innerportion of said second sleeve, and further wherein said first and secondsets of threads are of a first pitch, and third and fourth sets ofthreads are of a second pitch that is different than said first pitch.2. A temperature-compensating dielectric resonator tuning devicecomprising: a tuning element including a first portion thereof forelectromagnetically interacting with a dielectric resonator, whereinsaid tuning element is an elongated shaft and said first portion is saidend of said shaft; a first material disposed on said tuning element forcausing said portion of said tuning element to move with respect to saiddielectric resonator as temperature varies in accordance with a firstlinear temperature expansion coefficient, wherein said first material isconfigured into a first sleeve disposed coaxially around said shaft; anda second material disposed on said first material for causing saidportion of said tuning element to move with respect to said dielectricresonator as temperature varies in accordance with a second lineartemperature expansion coefficient, wherein said second material isconfigured into a second sleeve disposed coaxially around said firstsleeve; wherein said shaft includes a first set of threads formed aroundsaid shaft and configured to mate with a second set of threads formed onan inner portion of said first sleeve, and wherein said first sleeveincludes a third set of threads formed on an outer portion thereof, andconfigured to mate with a fourth set of threads formed on an innerportion of said second sleeve, and further wherein said second sleeveincludes a fifth set of threads for mating with a sixth set of threadsformed on an inner portion of a third sleeve disposed coaxially aroundsaid second sleeve.
 3. A dielectric resonator apparatus comprising: ahousing configured to form a cavity; a dielectric resonator situatedwithin said cavity; and a tuning device, comprising: a tuning elementincluding a first portion thereof for electromagnetically interactingwith a dielectric resonator, wherein said tuning element is an elongatedshaft and said first portion is said end of said shaft; a first materialdisposed on said tuning element for causing said portion of said tuningelement to move with respect to said dielectric resonator as temperaturevaries in accordance with a first linear temperature expansioncoefficient, wherein said first material is configured into a firstsleeve disposed coaxially around said shaft; and a second materialdisposed on said first material for causing said portion of said tuningelement to move with respect to said dielectric resonator as temperaturevaries in accordance with a second linear temperature expansioncoefficient, wherein said second material is configured into a secondsleeve disposed coaxially around said first sleeve; wherein said shaftincludes a first set of threads formed around said shaft and configuredto mate with a second set of threads formed on an inner portion of saidfirst sleeve, and wherein said first sleeve includes a third set ofthreads formed on an outer portion thereof, and configured to mate witha fourth set of threads formed on an inner portion of said secondsleeve, and further wherein said first and second sets of threads are ofa first pitch, and third and fourth sets of threads are of a secondpitch that is different than said first pitch.
 4. A dielectric resonatorapparatus comprising: a housing configured to form a cavity; adielectric resonator situated within said cavity; and a tuning device,comprising: a tuning element including a first portion thereof forelectromagnetically interacting with a dielectric resonator, whereinsaid tuning element is an elongated shaft and said first portion is saidend of said shaft; a first material disposed on said tuning element forcausing said portion of said tuning element to move with respect to saiddielectric resonator as temperature varies in accordance with a firstlinear temperature expansion coefficient, wherein said first material isconfigured into a first sleeve disposed coaxially around said shaft; anda second material disposed on said first material for causing saidportion of said tuning element to move with respect to said dielectricresonator as temperature varies in accordance with a second lineartemperature expansion coefficient, wherein said second material isconfigured into a second sleeve disposed coaxially around said firstsleeve; wherein said shaft includes a first set of threads formed aroundsaid shaft and configured to mate with a second set of threads formed onan inner portion of said first sleeve, and wherein said first sleeveincludes a third set of threads formed on an outer portion thereof, andconfigured to mate with a fourth set of threads formed on an innerportion of said second sleeve, and further wherein said second sleeveincludes a fifth set of threads for mating with a sixth set of threadsformed on an inner portion of a third sleeve disposed coaxially aroundsaid second sleeve.